Power factor correction circuits

ABSTRACT

A bi-directional boost circuit for power factor correction includes a power factor control circuit and a pair of diodes, a pair of inductors, and a pair of switches. A first diode, a second diode, a first inductor, a second inductor, a first switch, and a second switch convert the AC input voltage, rectify the AC input voltage, and output an intermediate DC voltage. The power factor control circuit receives the AC input voltage and receives the intermediate DC voltage. The power factor control circuit regulates the DC output voltage. Based on the AC input voltage and the intermediate DC output voltage, the power factor control circuit controls an inductor current waveform by driving the first switch and the second switch to create a substantially sinusoidal current as seen by the power source that is in phase with the AC input voltage.

BACKGROUND OF THE INVENTION

1. Technical Field

This invention relates to power adapters, and more specifically, to asystem and method to correct power factor, i.e., the ratio of real powerto apparent power.

2. Description of the Related Arts

The explosive growth in consumer electronics is causing the electricitysupply industry considerable concern. The appliances or consumerelectronics devices employ power supplies that draw current from the ACpower line during the peak of the sine wave. Most of the appliances orconsumer electronics device utilize a rectifier-bridge/smoothingcapacitor circuit.

Power factor is the ratio of real power to apparent power. In the UnitedStates, power is provided at approximately 120 Volts AC with a frequencyof approximately 60 Hertz. In Europe and other areas, power is providedat approximately 240 Volts AC with a frequency of approximately 50Hertz. In order to provide a maximum amount of usable energy or power,it is desirable for a load to draw current as if the load is entirelyresistive. If the load appears resistive, then the current drawn fromthe source may have a substantially sinusoidal shape, as the AC voltagehas, and the current drawn from the source may be in phase with the ACinput voltage.

Power supplies that utilize rectifier-bridge/smoothing capacitorcircuits draw non-sinusoidal currents as the AC line's instantaneousvoltage exceeds the storage capacitor's voltage. The electricitygenerator, with no power factor correction, must supply energy at thetop/peak of the sine wave rather than throughout the cycle, which cancause the sine wave to collapse around its peak.

The electricity generator sees the phase lag between the current andvoltage, together with the harmonics from peaky loads, as combining toprovide require rms currents, which in turn reduces the real power thatthe network can supply. Varying loads at the consumer end of the lineproduces fluctuations throughout the local line and these fluctuationscause undersirable consequences, such as causing lighting sources toflicker.

FIG. 1 illustrates the current and voltage waveforms for an electronicdevice that power factor correction (PFC) is designed to correctaccording to the prior art. As illustrated, the voltage waveform issinusoidal in shape and the current waveform can be characterized as awaveform with a steady current value with large spikes in the amplitudeof the current waveform along with a high content of harmonics. Thelarge spikes in the current waveform are caused because of the switchingpower supplies' use of the rectifier bridge/smoothing capacitorcircuits. From an efficiency viewpoint, a typical uncorrectedswitched-mode power supply has a power factor of 0.6, which effectivelyreduces the current available from the AC socket from about 13 to about7.8 Amps.

A solution for power factor correction is to condition the equipment'sinput load power so that it appears purely resistive using active PFCtechniques. Common PFC designs employ a boost preconverter ahead of theconventional voltage-regulation stage, which effectively cascades toswitched-mode power supplies. The boost preconverter raises thefull-wave rectified, unfiltered AC line to a DC input rail at a levelslightly above the rectified AC line, which is typically around 375 to400 volts DC. By drawing current throughout the AC line cycle, the boostpreconverter forces the load to draw current in phase with AC linevoltage, quashing harmonic emissions.

FIG. 2 illustrates a power factor correction circuit with a boostpreconverter according to the prior art. The full-wave bridge rectifier200 receives the AC input voltage and produces a full-wave rectifiedvoltage. The boost preconverter 205 receives the full-wave rectifiedvoltage and forces the load to draw current in phase with the voltage.The shape of the current waveform is determined by a switching device215, which is coupled to the output and a control circuit 220. Thecontrol circuit 220 provides an input to the switching device 215 andreceives as input signals a signal from the output and a signal from therectifier/boost node 225. This circuit may solve the power factorproblem by shaping the current waveform to mimic the voltage waveformand to cause the current waveform to be in phase with the voltagewaveform. However, the circuit utilizes at least five diodes, four ofwhich are located in the bridge rectifier, and diodes are lossycomponents, which decreases the power efficiency of the circuit.

Accordingly, it would be beneficial to have fewer lossy components in apower factor correction circuit, where the power factor correctioncircuit accepts a wide range of input voltages and automatically adjuststhe current waveform provided to be substantially sinusoidal in shapeand in phase with the AC input voltage waveform.

It would also be beneficial to utilize the circuitry that is rectifyingthe AC input voltage to assist in providing power factor correction.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a current and voltage waveforms for an electronicdevice that power factor conversion is designed to correct according tothe prior art;

FIG. 2 illustrates a power factor correction circuit with a boostpreconverter according to the prior art;

FIG. 3 illustrates a bi-directional boost circuit for power factorcorrection in a power adapter according to an embodiment of the presentinvention;

FIG. 3( a) illustrates programming resistors within a cable according toan embodiment of the present invention;

FIG. 3( b) illustrates programming resistors with a connector accordingto an embodiment of the present invention;

FIG. 4 illustrates a bi-directional boost circuit for power factorcorrection according to an embodiment of the present invention;

FIG. 5( a) illustrates a haversign signal according to an embodiment ofthe present invention.

FIG. 5( b) illustrates a ramp signal and a DC error signal according toan embodiment of the present invention;

FIG. 5( c) illustrates a pulsed signal according to an embodiment of thepresent invention;

FIG. 6( a) illustrates the inductor current output, as seen by the inputpower source, including the high frequency pulses in the substantiallysinusoidal envelope, according to an embodiment of the presentinvention;

FIG. 6( b) illustrates a clipped inductor current waveform as seen bythe power source according to an embodiment of the present invention;

FIG. 7 illustrates a schematic of a power factor correction circuitaccording to an embodiment of the present invention;

FIG. 8( a) illustrates a waveform created by a first waveform generatoraccording to an embodiment of the present invention;

FIG. 8( b) illustrates a waveform created by a second waveform generatoraccording to an embodiment of the present invention;

FIG. 8( c) illustrates a combination waveform generated by a firstwaveform generator and a second waveform generator according to anembodiment of the present invention;

FIG. 9( a) illustrates a second embodiment of the power factorcorrection circuit according to an embodiment of the present invention;

FIG. 9( b) illustrates current flow during the positive cycle of the ACinput according to an embodiment of the present invention; and

FIG. 9( c) illustrates a current flow during the negative cycle of theAC input according to an embodiment of the present invention.

DETAILED DESCRIPTION

FIG. 3 illustrates a bi-directional boost circuit for power factorcorrection according to an embodiment of the present invention. Theboost power factor conversion circuit may be utilized in an AC to DCpower adapter or power converter. It is desirable for the power adapterto appear to the power supply line as a resistive load. In other words,the current waveform should be in phase and have approximately the sameshape as the voltage waveform. This minimizes the demand on the utilityprovider because large, potentially instantaneous spikes of currentdemand may be avoided.

The power adapter 300 may include a pair of AC input terminals 301 and302, an electronic magnetic interference (EMI) filter 303, abi-directional boost circuit 304, a switching device 306, a transformer308, a regulator 310, and an error correction system 312. The poweradapter 300 may produce a power output having a regulated output voltageand a controlled output current.

The EMI filter 303 removes high-frequency noise from the power adapter.The bi-directional boost circuit 304 produces an intermediate DCvoltage. In addition, the bi-directional boost circuit 304 providespower factor correction for the inductor current, as seen by the powersource, and reduces any instantaneous current demand spikes. The poweradapter 300 may achieve power factor correction by utilizing thebi-directional boost circuit 304 to generate a current waveform, as seenfrom the power source, that is substantially sinusoidal in shape and inphase with the AC input voltage. The bi-directional boost circuit 304includes a control circuit 305 and a rectifying correction circuit orboost/switch circuitry 307. The rectifying correction circuit 307receives the AC input voltage and produces an intermediate DC outputvoltage by converting and rectifying the AC input voltage. The controlcircuit 305 receives the intermediate DC output voltage and the AC inputvoltage. Based on the intermediate DC output voltage and the AC inputvoltage, the control circuit 305 transmits a signal to cause therectifying correction circuit 307 to generate a current waveform, asseen by the power source, that is substantially sinusoidal in shape andin phase with the AC input voltage. The bi-direction boost circuit 304only utilizes two diodes so the number of lossy components is reducedand the power adapter efficiency is improved. Additionally, thebi-directional boost circuit 304 utilizes the same circuitry to boostand rectify the AC input voltage and also to generate a current waveformthat is substantially sinusoidal in shape and is in phase with the ACinput voltage. The current waveform may be an inductor current waveformor the current waveform as measured at the inductor in the boost/switchcircuitry 307.

The intermediate DC voltage is input to the switching device 306, whichoutputs a switched output. The switched output is input to a transformer308, which outputs a second intermediate voltage. The secondintermediate voltage is output to a regulator 310, which generates apower output with a regulated voltage and a controlled current. Thepower output is provided to a portable appliance 311. Because the powerrequirements of the portable appliances vary, e.g, CD players, need oneinput voltage and cell phones a second input voltage, the power outputof the power adapter may be regulated, as described below. Generally, aprogramming signal, i.e., a voltage programming signal or a currentprogramming signal, may be provided to an error correction system 312and the error correction system 312 can transmit a signal to theregulator 310 to regulate the output voltage or to control the outputcurrent.

Specifically, the regulator 310 may receive the second intermediatevoltage. In an embodiment of the invention, the regulator 310 may be abuck regulator, a boost regulator, or a buck-boost regulator, or anyother regulator commonly utilized in the power adapter art. Theregulator 310 generates a power output including a regulated voltage anda controlled current. The power output is provided to the portableappliance. Because different portable appliances have differentoperating voltage requirements and current needs, the power adapter 300may need to be configured to output the necessary regulated voltageand/or controlled current.

The regulated voltage and the controlled current output from theregulator 310 is also input to an error correction system 312. The errorcorrection system 312 may also receive a programming signal. Themagnitude of the programming signal may be dependent upon a value of aresistor located in a cable coupled to the power supply. In anembodiment of the invention, the magnitude of the programming signal maybe dependent upon a value of a resistor located in a connector coupledto the cable and the power supply. FIG. 3( a) illustrates a cable withprogramming resistors coupled to the power adapter and FIG. 3( b)illustrates a connector with programming resistors coupled to a cableand a power adapter according to an embodiment of the invention. In anembodiment of the invention, one or more resistors may be installed inone of the cable and the connector. The installation of the one or moreresistors in the one of the cable and the connector and the coupling ofthe cable or the cable and the connector, may change the magnitude ofthe programming signal to be transmitted to the error correction system312.

Illustratively, the one or more resistors may be coupled between a linein the cable or connector coupled to voltage program input and a line inthe cable or connector coupled to ground. Illustratively, the one ormore resistors may be coupled between a line in the cable or connectorcoupled to current program input and a line in the cable or connectorcoupled to ground. Illustratively, a first resistor may be coupledbetween a reference output and ground and a second resistor may becoupled between voltage program input and ground. This may produce adifferent magnitude of the programming signal because the first resistormay be coupled in parallel with a pullup resistor in the power adapter,which may modify the resistance value. Description of the resistorswithin the cable and the connector are found in the following patents,the disclosures of which are incorporated herein by reference: U.S. Pat.No. 5,838,554, entitled “Improved Small Form Factor Power Supply;” U.S.Pat. No. 5,949,213, entitled “Method and System for RechargingBatteries;” U.S. Pat. No. 6,172,884, entitled “Programmable PowerSupply;” and U.S. Pat. No. 6,266,261, entitled “DC Power AdapterSystem.”

Alternatively, a programming signal may be transmitted from an activedevice in the cable or from an active device in the connector. Theactive device may be a controller or an operational amplifier. Theactive device may transmit the programming signal to the errorcorrection system 312. The active device may receive a voltage referencesignal from the power adapter 300. Further discussion of such activedevices are provided in the following patent applications, thedisclosures of which are incorporated by reference: U.S. patentapplication Ser. No. 10/313,662, filed May 30, 2003, entitled “ActiveTip”, and U.S. patent application Ser. No. 10/313,793, filed Dec. 6,2002, entitled “Programmable Tip.”

Specifically, FIG. 4 illustrates an example of a bi-directional boostcircuit for power factor correction according to an embodiment of thepresent invention. In FIG. 4, the bi-directional boost circuit 307 ofFIG. 3 is presented in more detail. The bi-directional boost circuit mayinclude a first terminal 501, a second terminal 502, a first inductor503, a second inductor 504, a first diode 506, a second diode 508, anoutput terminal 510, a first switch 512, a second switch 514, and acapacitor 520. The control circuit 305 of FIG. 3 (control circuit 516 inFIG. 4) may include an error amplifier 531, a pulse width modulator 530,a waveform generator 532, an integrator 534, and a pair of switchcontrol circuits 536 and 538.

The first terminal 501 may be coupled to the first inductor 503, whichmay be coupled to the first diode 506 and a terminal of the first switch512. The second terminal 502 may be coupled to the second inductor 504,which may be coupled to the second diode 508 and a terminal of thesecond switch 514. The first diode 506 and the second diode 508 may becoupled to the output terminal 510, along with a terminal of thecapacitor 520, the other terminal of the capacitor 520 being coupled toground. The control circuit 516 may be coupled to the output terminal510, the first terminal 501, and the second terminal 502. The controlcircuit 516 may be coupled to the first switch 512 and the second switch514. In an embodiment of the invention, the control circuit 516 may becoupled to a control terminal of the first switch 512 and a controlterminal of the second switch 514, e.g., gate terminals when the firstswitch 512 and the second switch 514 are Field Effect Transistors.

The first terminal 501 and the second terminal 502 provides the AC inputvoltage from an input power source. The first inductor 503, the secondinductor 504, first diode 506, the second diode 508, the first switch512, and the second switch 514 receive the AC input voltage, convertsthe AC input voltage, rectifies the AC input voltage, and produces anintermediate DC output voltage, as discussed further below.

The first switch 512 may be a field effect transistor (FET) that has afirst terminal coupled to the junction between the first inductor 503and the first diode 506. The second switch 514 may be a FET and have afirst terminal coupled to the junction between the second inductor 504and the second diode 508. A second terminal of the first switch 512 anda second terminal of the second switch 514 may be coupled to ground.

During the positive cycle of the AC input, i.e., if a voltage atterminal 501 is greater than a voltage at a terminal 502, and the firstswitch 512 is on, energy is stored in the first inductor 503. If thefirst switch 512 is off, the energy stored in the first inductor istransferred through the first diode 506 to the capacitor 520. Thiscauses the capacitor 520 to charge to a steady state DC voltage Vposduring the positive cycle of the AC input. The combination of the firstdiode 506 and the first switch 512 are utilized to rectify the AC inputin the positive cycle. In an embodiment of the invention, the capacitor520 may also smooth the ripple of the rectified DC output voltage.

During the negative cycle of the AC input, e.g., if a voltage atterminal 502 is greater than a voltage at terminal 501, and the secondswitch 514 is on, energy is stored in the second inductor 504. If thesecond switch 514 is not on, the energy stored in the second inductor504 is transferred through the second diode 508 to charge up thecapacitor 520 to a steady state voltage Vpos2. In an embodiment of theinvention, the switches 512 and 514 may have an operational frequency of80 to 120 Kilohertz. Although the magnitude of the AC input is negativeduring the negative cycle of the AC input, the current is flowing to thecapacitor 520 in the same direction as the current flowing during thepositive cycle of the AC input, and thus the voltage across thecapacitor 520 is positive. The combination of the second diode 508 andthe second switch 514 are utilized to rectify the AC waveform. In anembodiment of the invention, the capacitor 520 may also smooth theripple of the rectified DC output. The intermediate DC output voltage isthe addition of steady-state voltages Vpos and Vpos2, and has the shapeof a rectified waveform, as smoothed by the filtering action of thecapacitor 520.

The AC input voltage may also be provided to the power factor controlcircuit 516. In addition, the power factor control circuit 516 mayreceive the intermediate DC output voltage from the output terminal 510.The control circuit 516 may utilize the AC input and the DC output tocreate driving signals that are respectively input to the control, e.g.,gate, terminals of the first switching device 512 and the secondswitching device 514. The control circuit 516 may control the current inthe inductors 503 and 504 and may cause an inductor current to have asubstantially sinusoidal shape. The substantially sinusoidal shape ofthe inductor current corresponds to a haversign signal generated withinthe control circuit 516 except that the inductor current sinusoidalwaveform crosses a line, e.g., a reference potential while the haversignsignal waveform does not cross a line on a graph and stays positive, theline, for example, representing a reference potential.

Specifically, the AC input voltage may be input to a waveform generator532. The waveform generator 532 may generate a haversign waveform, asillustrated in FIG. 5( a). The intermediate DC output voltage is inputto an error amplifier 531. The intermediate DC output voltage iscompared to a reference voltage and a DC error signal is output from theerror amplifier 531. A pulse width modulator 530 receives the DC errorsignal from the error amplifier 531 and compares the DC error signal toa ramp signal. In an embodiment of the present invention, the rampsignal may be oscillating in a frequency range of 80-120 Kilohertz. TheDC error signal is illustrated as a dotted line in FIG. 5( b) and theramp signal is illustrated as a solid line in FIG. 5( b). The pulsewidth modulator 530 outputs a pulsed signal. Illustratively, the pulsedsignal for the comparison of the DC error signal and the ramp signal isillustrated in FIG. 5( c).

The haversign waveform and the pulsed signal are multiplied together tocreate a multiplied haversign waveform, which is input to an integrator534. The amplitude of the multiplied haversign waveform is controlled bythe pulsed signal. The integrator 534 strips off the high frequencycharacteristics of the multiplied haversign waveform and produces ahaversign signal.

In an embodiment of the invention, a haversign signal may be input to afirst switch control circuit 536 and a second switch control circuit538. During the positive cycle of the AC input, the first switch controlcircuit 536 monitors the actual current in the first inductor 503. Thefirst switch control circuit compares a magnitude of the inductorcurrent in the first inductor 503 to a threshold value, e.g., amagnitude of the haversign signal at specific point in time. When theinductor current is greater than the magnitude of the haversign signalinput to the first switch control circuit 536, the first switch controlcircuit transmits a first drive signal to turn off the first switch 512.Because the haversign signal is sinusoidal in shape, the inductorcurrent output during the positive cycle of the AC input from the firstswitch 512 may also have a sinusoidal shape. The frequency of a signaloutput from the first switch control circuit 536 is a high frequency,e.g., in the range of 80 Kilohertz to 120 Kilohertz. Because the firstswitch 512 is turning on and off at the high frequency, the inductorcurrent output is actually a series of high frequency pulses operatingwhich are formed in a substantially sinusoidal envelope, where thesinusoidal envelope is oscillating between 45-65 Hertz. FIG. 6( a)illustrates the inductor current output, as seen by the input powersource, including the high frequency pulses in the substantiallysinusoidal envelope. This may be referred to as an inductor currentincluding a substantially sinusoidal shape. Due to the high frequency atwhich the first switch 512 operates, FIG. 6( a) is not drawn to scale.

During the positive portion of the input AC waveform, the second switch514 is always on because a threshold value is never reached, i.e., theinductor current measured in the second inductor 504 by the secondswitch control circuit 538 never reaches the value of the haversignsignal input to the second switch control circuit 538. Therefore, thesecond switch 514 is always turned on during the positive cycle of theAC waveform.

The haversign signal is also input to the second switch control circuit538 and the second switch control circuit 538 outputs a second drivesignal to drive the second switching device 514 during the negativecycle of the AC input. Again, because the haversign waveform issinusoidal in shape, the inductor current, as seen from the powersource, may also be sinusoidal in shape, i.e., may be a number of highfrequency pulses that ride in a substantially sinusoidal envelope.During the negative cycle of the AC input, the magnitude of thissubstantially sinusoidal waveform may be negative, unlike the haversignsignal which has a positive magnitude.

The power source may see the inductor current waveforms during thepositive cycle and the negative cycle of the AC input as a substantiallysinusoidal waveform that is in phase with the AC input voltage. Powerfactor correction is achieved because the substantially sinusoidalwaveform is resistive in nature and the instantaneous peak demands forcurrent has been reduced or eliminated.

In an embodiment of the invention, the first switch control circuit 536and the second switch control circuit 538 may limit the amplitude of theinductor current. This may be beneficial for use in the United States,where power factor correction is not required, and thus the inductorcurrent may not need to be sinusoidal in shape. In order to output thenecessary power within the United States, approximately double theamount of current may need to be utilized in order to produce the samepower with half the current in Europe. This large amount of current mayrequire larger size inductors to handle the amount of current in thedevice which may result in less efficiency of the power supply. Thisloss of efficiency may be neutralized by clipping the substantiallysinusoidal waveform of the inductor current as seen from the powersource, i.e., limiting the peak current of the sinusoidal waveform, andproducing more of a trapezoidal- or square-shaped inductor currentwaveform.

In this embodiment of the invention, the first switch control circuit536 may limit the amplitude of the inductor current on the positive halfof the AC input if the magnitude of the inductor current exceeds aclipping threshold value. In other words, the first switch controlcircuit 536 clips the peak of the inductor current waveform, making thewaveform have more of a square-wave or a trapezoidal shape as seen bythe power source, rather than a sinusoidal shape. In this embodiment,the peak current in the inductors is lower. During the negative cycle ofthe waveform, the second switch control circuit 538 may also limit theamplitude of the inductor current if the magnitude exceeds the clippingthreshold value. FIG. 6( b) illustrates an ideal version of a clippedinductor current waveform as seen by the power source according to anembodiment of the present invention. Again, the actual inductor currentwould be a series of high frequency pulses which are either square ortrapezoidal in shape. A clipping threshold value may be set byinstalling a component across pins of the first switch control circuit536 or the second switch control circuit 538. FIG. 6( b) illustrates thehigh frequency pulses that create the clipped inductor current waveform.Again, FIG. 6( b) is not drawn to scale due the high frequency of theswitches being activated.

FIG. 7 illustrates a schematic of the power factor correction circuitaccording to an embodiment of the present invention. A first terminal601 of the AC line is coupled to a first terminal of a first inductor604 and a waveform generator 670 (through AC-A). A second terminal 602of the AC line is coupled to a first terminal of a second inductor 605and the waveform generator 670 (through AC-B). A second terminal of thefirst inductor 604 is coupled to a first diode 610 anode and is coupledan output terminal of a first switching device 614. A second terminal ofthe second inductor 605 is coupled to a second diode 612 anode and anoutput terminal of a second switching device 616.

The control terminal of the first switching device 614 is coupled to thefirst current mode controller 617. The control terminal of the secondswitching device 616 is coupled to the second current mode controller618. The first diode 610 cathode and the second diode 612 cathode arecoupled to the output terminal 640 of the power factor correctioncircuit 600. The output terminal 640 is coupled to a first erroramplifier 620 and also may be coupled to a second amplifier 652.

The error amplifier 620 compares a reference voltage with the highvoltage intermediate DC output present at the output terminal 640. Theerror amplifier 620 outputs a voltage error signal. For example, in theembodiment of the invention illustrated in FIG. 7, the voltage input tothe inverting input of the error amplifier 620 may be approximately 5.0volts, which is determined after the high voltage DC output is input toa voltage divider network created by resistors R26 and R27. In theembodiment of the invention illustrated in FIG. 7, the voltage input tothe inverting input of the error amplifier 620 may be compared to areference voltage of 5.1 volts.

The voltage error signal output from the error amplifier 620 may beinput into a pulse width modulator 630. The pulse width modulator 630may compare a periodic ramp signal with the DC signal output from theerror amplifier 620. The pulse wide modulator 630 may generate a pulsedsignal. In an embodiment of the invention, the ramp signal may begenerated by the current mode controllers 617 and 618. For example, thevalues of the resistor R17 and capacitor C6 attached to pin 4 of thecurrent mode controller 618 may determine the frequency of the rampsignal, as illustrated by the box marked Ramp in FIG. 7.

The pulsed signal may be multiplied by a haversign signal generated by awaveform generator 670. The waveform generator 670 may include a firstwaveform generator 671 and a second waveform generator 672. FIG. 8( a)illustrates a waveform created by a first waveform generator accordingto an embodiment of the invention. A first waveform generator 671receives the AC input from terminals 601 and 602 and acts as adifferential amplifier to create a sinusoidal shaped waveform for apositive cycle of the AC input and no waveform for the negative cycle ofthe AC input, as illustrated in FIG. 8( a). A second waveform generator672 receives the AC input from terminals 601 and 602, except the secondwaveform generator subsystem has the signal from terminal 601 input tothe non-inverting input of the operational amplifier U4B and the signalfrom terminal 602 input into the inverting input of the operationamplifier U4B, which is opposite to the inputs to the first waveformgenerator. FIG. 8( b) illustrates a waveform created by a secondwaveform generator according to an embodiment of the present invention.The second waveform generator 672 creates a sinusoidal waveform oppositein phase to the waveform generated by the first waveform generatorsubsystem 671, as illustrated in FIG. 8( b), i.e., a sinusoidal waveformis generated during the negative cycle of the AC input and no waveformis generated during the positive cycle of the AC input.

The output of the first waveform generator 671 and the second waveformgenerator 672 are summed at node 675. The resulting waveform is ahaversign waveform, as illustrated in FIG. 8( c).

The multiplication of the haversign and the pulsed signal may produce amultiplied haversign output at node 676. In other words, the pulsedsignal may control the magnitude of the haversign.

The integrator 650 strips the multiplied haversign output of highfrequency characteristics created by the pulse wide modulator's 630operating frequency and creates an integrated haversign waveform. Thehigh frequency characteristics are generated by the pulse widthmodulator 630 pulsing at the ramp signal frequency. In one embodiment ofthe invention, the ramp signal frequency may be 100 Kilohertz. Asillustrated in FIG. 7, the integrator 650 may be formed by a resistorR15 and a capacitor C19.

Comparator U5A 652 may respond to instantaneous large changes in themagnitude of the intermediate DC output voltage. Comparator U5A 652 maygenerate a clamp signal to prevent the control circuit from respondingto large instantaneous changes in intermediate DC output voltagemagnitude. If the large instantaneous change in the magnitude of theintermediate DC output voltage is detected by U5A 652, U5A 652 mayoutput a signal to clamp the integrated haversign waveform.

U4C 654 is a buffer for the integrated haversign waveform and outputs ahaversign signal. The integrated haversign waveform is input to thenon-inverting input of the amplifier and the feedback signal from U4C654 is input to the inverting input of U4C 654. These connections enablethe operational amplifier U4C 654 to have a unity gain at the output andto operate as a buffer.

A control circuit 680 controls current in the inductors L2 (L2-A andL2-B) 604 and L3 (L3-A and L3-B) 605. The current in the inductors 604and 605 is controlled by having the current track the voltage output,i.e., the haversign signal, from U4C 654. The current mode controllers617 and 618 control the current flow through the inductors 604 and 605by turning off and on switches 614 and 616, respectively. The input tothe current mode controllers 617 and 618 is a voltage from U4C with awaveshape of a haversign, i.e., a haversign signal. For example, thecurrent mode controllers 617 and 618 are voltage-to-current convertersso the inductor current tracks the waveform shape of the haversignsignal input to the current mode controllers 617 and 618. When the ACinput reverses, i.e., goes to the negative cycle, the inductor currentis traveling in the opposite direction, i.e., has a negative value, andinstead of precisely tracking the haversign signal, the inductor currentwaveform maintains the shape of the haversign signal, but the inductorcurrent is negative with respect to a reference potential and theinductor current crosses the reference potential when the AC inputreverses. Therefore, the resulting AC waveform is a substantiallysinusoidal current that crosses a reference potential when it switchesfrom a positive cycle to a negative cycle.

The current mode controllers 617 and 618 receive the haversign signalfrom the buffer U4C 654. In the embodiment of the invention illustratedin FIG. 6, the haversign signal is input to pins 1 and 2 of current modecontrollers 617 and 618. During the positive cycle of AC input, e.g.,when the voltage at terminal 601 is greater than the voltage at terminal602, the current mode controller 617 monitors the inductor currentthrough inductors L2 604. In an embodiment of the invention, the currentmode controller 617 monitors the inductor current via pin 3 by measuringthe current across resistor R22. The current mode controller 617compares the monitored inductor current to a threshold value.Illustratively, the threshold value is the input from buffer U4C 654,which is a value of the haversign signal at an instant in time. If themonitored inductor current is larger than the threshold value, then thecurrent mode controller 617 may turn off the switching device 614. Inother words, the current mode controller 617 is allowing the inductorcurrent to track up the value of the haversign signal at that moment intime, but does not allow the inductor current to go higher than thevalue of the haversign signal. Thus, the current in the inductor 604tracks the haversign waveform.

The current mode controller 618 activates switch 616 continuously duringthe positive cycle of the AC input. The current mode controller 618monitors the current in the inductors 605 and compares it to thethreshold value. The inductor current being monitored has a negativevalue, however, because the current is flowing from common groundthrough the switch 616 and back to the inductor 605. Because thethreshold value is positive, the current value never reaches thethreshold value and switch 616 is continuously activated during thepositive cycle of the AC input. For example, during the positive cycleof the AC input, the current path is from inductors 604 through switch614 through resistor R22 to common ground. Then, the current pathreturns from common ground through resistor R29 through switch 616 andto inductors 605. The current mode controller 618 is monitoring theinductor current, which is flowing in a negative direction through theresistor which is utilized for monitoring, e.g., R19. Because the valueof the inductor current is negative, the value of the inductor currentnever reaches the threshold value, and switch 616 is always turned onduring the positive cycle of the AC input.

Conversely, when the AC input is in a negative cycle, the value of theAC input at terminal 602 is greater than the value of the AC input atterminal 601, current mode controller 618 turns on and off switch 616and current mode controller 617 turns switch 614 on continuously.Illustratively, during the negative cycle of the AC input, the currentmode controller 618 is monitoring the inductor current in inductors 605when switch 616 is on via resistor R19. The current mode controller 618compares the inductor current to the value of the haversign signal inputfrom buffer U4C 654. If the inductor current is larger than the voltageinput, then the switch 616 is turned off. Thus, the current in theinductors 605 tracks the shape of the input voltage from amplifier 654,i.e., the haversign shape. Because the AC input is in the negative cycleand the value is negative with respect to a reference potential, theinductor current tracks the haversign shape but has a negative valuewith respect to a reference potential. Thus, during the positive andnegative cycle of the AC input, the inductor current waveform issinusoidal and crosses the reference potential when the AC input movesfrom the positive cycle to the negative cycle, and vice versa.

When the AC input is in a negative cycle, current mode controller 617activates switch 614 continuously. The current path when the AC input isin a negative cycle is for inductor 605 through switch 616 to resistorR19 to common ground, back from common ground through R22 through switch614 and back to inductor 604. Thus, the value of inductor current acrossR22 is negative. The value of the inductor current is monitored bycurrent mode controller 617. Because the value is negative, the value ofthe inductor current never reaches the threshold value established bythe haversign signal input to the current mode controller 617 from U4C654. Thus, the current mode controller 617 does not turn switch 614 offduring the negative cycle of the AC input.

As discussed previously, in the United States, power factor correctionis not necessary. Therefore, the current waveform may not need to besinusoidal in shape. If the power adapter is to be utilized within theUnited States, the current mode controllers 617 and 618 may beconfigured to limit the peak value of the inductor current. The inductorcurrent may be limited by configuring the current mode controllers 617and 618 to utilize a lower threshold value, and not the threshold valueinput from buffer U4C 654. This may create a current waveform that has asquarewave or a trapezoidal waveform rather than a sinusoidal waveform.The power adapter may still deliver the necessary power to the portableappliance. The current mode controllers 617 and 618 may compare, duringthe positive and negative cycles of the AC input, respectively, theinductor current to a clipping threshold value. The clipping thresholdmay be the lower threshold value. If the inductor current is higher thanthe clipping threshold, then the current mode controller 617 and 618 mayturn off switches 614 and 616, respectively. This may result in acurrent waveform that has a squarewave shape or a trapezoidal shape.

The power factor correction circuit of the present invention has anincreased efficiency due to the lower number of lossy components, i.e.,diodes, that are utilized in the design. In a standard power factorcorrection circuit, at least five diodes are utilized (four of thediodes being utilized in a bridge rectifier). In the power factorcorrection circuit of the present invention, only two diodes areutilized.

In regards to the voltage, the power factor correction circuit 600 mayreceive the AC input voltage on the AC input first terminal 601 and theAC input second terminal 602. The AC voltage input may be rectified toproduce a rectified DC voltage. The AC input is rectified utilizing thefirst diode 611, the second diode 612, the diode characteristics of thefirst switching device 614, and the diode characteristics of the secondswitching device 616. In other words, two actual diodes, the first diode610 and the second diode 612, are utilized along with the diodecharacteristics of the first switching device 614 and the secondswitching device 616 to rectify the AC input voltage.

The intermediate DC output voltage is created during a positive cycle ofthe AC waveform, when switch 614 opens and the energy that has beenstored in inductors L2 604 is transferred through diode 610 to chargecapacitors C2 and C3 620. The current input to the capacitor 620 createsa rectified DC output for the positive cycle of the AC input. Thecurrent then is returned to terminal 602 via a path that includes goingthrough the reference ground to R19, switching device 616, and inductors605.

The intermediate DC output voltage is created during the negative cycleof the AC waveform, when switch 616 opens and the energy that has beenstored in inductors L3 605 is transferred through diode 612 to chargecapacitors C2 and C3 625. The current is flowing across the capacitorsC2 and C3 625 in the same direction as it is during the positive cycleof the AC, so the voltage across the capacitors C2 and C3 625 ispositive. The current path during the negative cycle of the AC waveformfollows the path of reference ground through R22, switching device 614,and inductors 605 to terminal 601.

The voltage waveform created during the positive cycle and the negativecycle is added to create an intermediate DC output voltage. Thecapacitors C2 and C3 625 filter the full-wave rectified voltage tocreate the intermediate DC output voltage with minimal ripple.

FIG. 9( a) illustrates a second embodiment of the power factorcorrection circuit according to an embodiment of the present invention.In this embodiment, the power factor correction circuit replaces theboost and switch circuitry 304, the switching device 306, and thetransformer 308 in FIG. 3. A control circuit 516, as described in FIG.4, may be utilized with the embodiment illustrated in FIG. 9( a). In anembodiment of the invention, a slightly modified control circuit 516 maybe utilized with the embodiment illustrated in FIG. 9( a). In oneembodiment of the invention, the power factor correction circuit 900 maybe located in AC to DC power converter. In an embodiment of theinvention the power factor correction circuit 900 may be located in anelectronic device that receives as input an AC voltage and needs tomaintain a waveform that meets with European power drawing requirements,e.g., such as a sinusoidal shaped current waveform. The power factorcorrection circuit of the present invention produces an isolatedpower-factored corrected rectified current waveform at the intermediatenode 930. In the illustrated embodiment, only two diodes are utilized inthe power factor correction circuit 900, which results in a lower powerloss and energy consumption for the circuit, which normally utilized atleast six diodes. The present invention handles power factor correction,isolating, transforming of the voltage, and rectifying of the voltage inone power stage, which eliminates the need for a bridge rectifiercircuit.

The power factor correction circuit 900 receives an AC input voltage viaa first AC input terminal 901 and a second AC input terminal 902. Thefirst AC input terminal 901 is coupled to a first primary winding 906 ofthe transformer 920. The first primary winding is coupled to a firstterminal of a first switch 904. A control terminal of the first switch904 is coupled to a control circuit 903. A second terminal of the firstswitch is coupled to a reference ground.

The second AC input terminal 902 is coupled to a first terminal of asecond primary winding 908. A second terminal of the second primarywinding is coupled to a first terminal of a second switch 905. A secondterminal of the second switch 905 is coupled to the reference ground. Acontrol terminal of the second switch 905 is coupled to a controlcircuit 903. In an embodiment of the invention, the control circuit 903may be configured so that both the first switch 904 and the secondswitch 905 turn on simultaneously. In an embodiment of the invention,the control circuit 903 may be configured so that one switch is alwaysturned on and the other switch may be turned off and on. For simplicityof discussion, the embodiment where the first switch 904 and the secondswitch 905 are turned on simultaneously is discussed below.

The transformer 920 includes a first primary winding 906, a secondprimary winding 908, a transformer core 910, a first secondary winding912, and a second secondary winding 914. In an embodiment of theinvention, the transformer 920 may utilize planar magnetics. Traditionalmagnetics with solid or stranded wire may also be used utilized, howeverthe effect of the noise cancellation outlined below may be reduced.

In an embodiment of the present invention, the physical configuration ofthe transformer 920 forms a pair of capacitors 940 and 942. The firstprimary winding 906 and the second primary winding 908 are both locatedon a circuit board that acts as a plate for the pair of capacitors. Thefirst secondary winding 912 and the second secondary winding 914 arelocated on a circuit board that acts as a second plate for the pair ofcapacitors. The core 910 is the dielectric material for the pair ofcapacitors. Capacitor 940 is formed between the first primary winding906 and the first secondary winding 912. Capacitor 942 is formed betweenthe second primary winding 908 and the second secondary winding 914.

A first terminal of the first secondary winding 912 is coupled to afirst terminal of a first rectification diode 916. A second terminal ofthe first secondary winding 912 is coupled to a reference ground. Afirst terminal of the second secondary winding 914 is coupled to thereference ground. A second terminal of the second secondary winding 914is coupled to a first terminal of a second rectification diode 918.

A second terminal of the first rectification diode 916 and the secondrectification diode 918 are coupled to an intermediate node 930. Theintermediate node 930 is coupled to a first terminal of a capacitor 925and a second terminal of the capacitor 925 is coupled to the referenceground. In an embodiment of the invention, MOSFETS may be utilized inplace of the rectification diodes 916 and 918 as active rectifiers, alsois known as synchronous rectification.

In operation, the power factor correction circuit provides a regulatedintermediate DC voltage at the intermediate node 930 and across thecapacitor 925. In an embodiment of the invention, the power factorcorrection circuit generates a substantially sinusoidal current waveformthat enables a power factor converter to meet the input line harmonicrequirements of EN691000-3-2. In an embodiment of the invention, thepower factor correction circuit generates a clipped current waveform. Inan embodiment of the invention, the waveform generated by the powerfactor correction circuit is seen or viewed from the input terminals,and thus the power supplier.

FIG. 9( b) illustrates current flow during the positive cycle of the ACinput according to an embodiment of the present invention. The positivecycle of the AC input is defined as when the voltage at terminal 901 isgreater that the voltage at terminal 902. During the positive cycle ofthe AC input, when the first switch 904 and the second switch 905 areon, the current flows from the first AC input terminal 901 to the firstterminal of the primary winding 906 to the second terminal of theprimary winding 906 through the first terminal of the switch 904 to thesecond terminal of the first switch and then to the reference ground.The current then flows from the second terminal of the second switch 905from the reference ground through the first terminal of the secondswitch 905 to the second terminal of the second primary inductor 908through the first terminal of the second primary inductor 908 and to thesecond input terminal 902.

As the first switch 904 and the second switch 905 are switched off, thecurrent is induced via the magnetic core to the secondary side of thetransformer 920. Due to the configuration of the transformer windings,which are indicated by the dots on the transformer 920, during thepositive cycle of the AC input, the current flows from the referenceground to the second terminal of the first secondary winding 912 to thefirst terminal of the first secondary winding 912 through the firstrectification diode 916 to the first intermediate node 930 to thecapacitor 925 and then to the reference ground.

In one embodiment of the invention, the current waveform may besubstantially sinusoidal in shape after exiting the first terminal ofthe first secondary winding 912. After exiting the first terminal of thefirst secondary winding, the current waveform may be a large number ofpulses forming a substantially sinusoidal envelope. The firstrectification diode 916 may rectify the current waveform to produce ahaversign waveform.

The voltage transferred to the intermediate node is dependent on theturns ratio of the primary windings to the secondary windings. Duringthe positive cycle of the AC input, when the first switch 904 and thesecond switch 905 are on, energy is stored up in the core 910 of thetransformer 920. During this time the voltage across the first primarywinding 906 may be labeled as V_(primary1). The voltage on the secondaryside, i.e., V_(secondary1) of the transformer is determined bymultiplying V_(primary1)*(N_(secondary)/N_(primary)) where N representsthe number of turns in the windings. For example, if the voltage acrossthe first primary winding is 200 volts, the turns ratio is 20 to 1, andthe voltage being regulated to at intermediate node 930 is, for example+15 volts, then when the first switch 904 and the second switch 905 areon, V_(secondary1) (the voltage across the first secondary winding) isequal to −10 Volts. V_(secondary1) is −10 Volts, that is the anode ofrectifying diode 916 is −10 volts with respect to the secondaryreference ground, because of the orientation of the first secondarywinding of the transformer 920. This results in the back-biasing of thefirst rectifying diode 916 and thus no voltage is transferred tointermediate node 930.

In this example, V_(secondary2) (the voltage across the second secondarywinding) is also −10 volts, that is, the anode of the second rectifyingdiode 918 is +10 volts with respect to secondary reference ground. Thevoltage does not move across the second rectifying diode 918 because theanode of the second rectifying diode 918 has +10 volts and the cathodeof the second rectifying diode has +15 volts so no voltage passes to theintermediate node 930 and the capacitor 925.

During the positive cycle of the AC input, when the first switch 904 andthe second switch 905 are turned off, the energy that was stored in thecore 910 of the transformer 920 is transferred into the windings on thesecondary side of the power factor correction circuit. V_(secondary1)continues to build up voltage until it is greater than V_(intermediate)(the voltage at the intermediate node 930) by the forward drop ofrectifying diode 916 and then V_(secondary1) transfers energy throughrectifying diode 916 to the intermediate node 930 and the capacitor 925to build up the V_(intermediate) to the desired regulating voltage.Illustratively, if the desired regulating voltage is 17 volts, then thecontrol circuit 903 receives this information from the power converter,drives the switches to generate a corresponding voltage on the primaryside of the transformer 920, and transfers this energy to the secondaryside of the transformer 920. Due to the orientation of the secondsecondary winding 914, V_(secondary2) is driven to a negative voltagewhen the first switch and the second switch are turned off band thus novoltage moves across the second rectifying diode 918.

FIG. 9( c) illustrates current flow during a negative cycle of the ACinput according to an embodiment of the present invention. During thenegative cycle of the AC input, when the voltage at input terminal 902is greater than the voltage at input terminal 901, the current flows,when the first switch 904 and the second switch 905 are on, asillustrated in FIG. 9( c). The current flows from the second inputterminal 902 through the first and second terminals of the secondprimary winding 908, the first and second terminals of the switch 905 toreference ground. The current then flows from the reference groundthrough the second terminal and the first terminal of the first switch,through the second terminal and the first terminal of the first primarywinding 906 to the first input terminal 901.

As the first switch 905 and the second switch 905 are switched off, thecurrent is induced via the magnetic core to the secondary side of thetransformer 920, specifically the second secondary winding 914. Due tothe configuration of the transformer windings, which are indicated bythe dots on the transformer 920, during the negative cycle of the ACinput, the current flows from the reference ground to the first terminalof the second secondary winding 914 to the second terminal of the secondsecondary winding 914 through the second rectification diode 918 to thefirst intermediate node 930 to the capacitor 925 and then to thereference ground.

In one embodiment of the invention, the current waveform may besubstantially sinusoidal in shape after exiting the first terminal ofthe second secondary winding 914. After exiting the second terminal ofthe second secondary winding, the current waveform may be a large numberof pulses forming a substantially sinusoidal envelope. The secondrectification diode 918 may rectify the current waveform to produce asecond half of the haversign waveform. The first rectifying diode 916and the second rectification diode rectify the current going into node930

During the negative cycle of the AC input, when the first switch 904 andthe second switch 905 are on, energy is stored up in the core 910 of thetransformer 920. V_(secondary2) is −10 Volts, that is the anode ofrectifying diode 918 is −10 volts with respect to the secondaryreference ground, because of the orientation of the second secondarywinding 914 of the transformer 920. This results in the back-biasing ofthe second rectifying diode 918 and no voltage is transferred tointermediate node 930. In this example, V_(secondary1) (the voltageacross the first secondary winding) is +10 volts, that is the anode ofrectifying diode is +10 volts with respect to the secondary referenceground. The voltage does not move across the first rectifying diode CR2because the anode of the first rectifying diode 916 has +10 volts andthe cathode of the second rectifying diode 918 has +15 volts (from theregulating voltage) reverse biasing the second rectifying diode 917 sono voltage passes to the intermediate node 930 and the capacitor 925.

During the negative cycle of the AC input, when the first switch 904 andthe second switch 905 are turned off, the energy that was stored in thecore 910 of the transformer 920 is now transferred into the windings onthe secondary side of the power factor correction circuit.V_(secondary2) builds up voltage until it is greater thanV_(intermediate) (the voltage at the intermediate node 930) by theforward drop of the rectifying diode 918, and then V_(secondary2)transfers energy through rectifying diode 918 to the intermediate node930 and the capacitor 925 to build up V_(intermediate) to the desiredregulating voltage. Due to the orientation of the first secondarywinding, V_(secondary1) is driven to a negative voltage when the secondswitch is turned off band thus no voltage moves across the firstrectifying diode 916.

The power factor correction circuit 900 also results in the cancellationof high frequency common mode noise generated by the high frequencyswitching of the first switch 904 and the second switch 905. Asdiscussed above, the first primary winding 906 is capacitively coupledto the first secondary winding 912 and the second primary winding 908 iscapacitively coupled to the second secondary winding 914. If both thefirst switch 904 and the second switch 905 are turned on, the voltageacross the first capacitive coupling (the first primary winding 906 andthe first secondary winding 912) and the voltage across the secondcapacitive coupling (the second primary winding 908 and the secondsecondary winding 914) are going to be equal in amplitude. The voltagesacross the capacitive couplings are going to be equal in amplitude, butopposite in phase. In other words they are going to be going in oppositedirections. The voltage across the first capacitive coupling is, forexample, +200 volts with respect to the reference ground; while thevoltage across the second capacitive coupling is, for example, −200volts. Thus, perfect high frequency common noise generation results. Inother words you have a capacitor divider where the net sum energy iszero. Illustratively, during the negative cycle of the AC input, avoltage across the capacitor 942 may be +100 volts which is appliedacross the second terminal and the first terminal of the secondsecondary winding 914. Also, during the negative cycle of the AC input,a voltage across capacitor 940 may be −100 volts, which is appliedacross the first terminal and the second terminal of the first secondarywinding 912. In this example, the voltages cancel each other outresulting in common mode noise cancellation.

While the description above refers to particular embodiments of thepresent invention, it will be understood that many modifications may bemade without departing from the spirit thereof. The accompanying claimsare intended to cover such modifications as would fall within the truescope and spirit of the present invention. The presently disclosedembodiments are therefore to be considered in all respects asillustrative and not restrictive, the scope of the invention beingindicated by the appended claims, rather than the foregoing description,and all changes which come within the meaning and range of equivalencyof the claims are therefore intended to be embraced therein.

1. A bi-directional boost circuit for power factor correction,comprising: a first diode, a second diode, a first inductor, a secondinductor, a first switch, and a second switch to convert an AC inputvoltage, rectify the AC input voltage, and output an intermediate DCvoltage; and a power factor control circuit, the power factor circuitincluding a waveform generator to receive the AC input voltage andgenerate a haversign waveform; a pulse width modulator to generate apulsed signal based on the intermediate DC voltage; a multiplier tomultiply the haversign waveform and the pulsed signal and to create amultiplied haversign signal; an integrator to strip off high frequencycharacteristics of the multiplied haversign signal to create a haversignsignal; a first control circuit to compare a magnitude of the haversignsignal to a magnitude of a first inductor current to generate a firstdrive signal; and a second control circuit to compare a magnitude of thehaversign signal to a magnitude of a second inductor current to generatea second drive signal, which controls an inductor current waveform toform a substantially sinusoidal waveform that is in phase with the ACinput voltage.
 2. The circuit of claim 1, wherein the power factorcontrol circuit creates a clipped inductor current waveform if the powerfactor control circuit is configured with a clipping threshold value anda value of an inductor current is above the clipping threshold value. 3.The circuit of claim 1, wherein the first control circuit is a firstcurrent mode controller and the second control circuit is a secondcurrent mode controller.
 4. The circuit of claim 1, further including anerror amplifier to compare a reference voltage and the intermediate DCvoltage and to generate an error signal and to input the error signal tothe pulse width modulator to control a pulse width of the pulsed signal.5. The circuit of claim 1, further including a comparator to clamp thehaversign signal in response to a large instantaneous change in theintermediate DC output voltage.
 6. A method of power factor correction,comprising: receiving an AC input voltage; generating an intermediate DCvoltage; generating, at a waveform generator, a haversign waveform;generating a pulsed signal based on the intermediate DC voltage;multiplying the haversign waveform and the pulsed signal to create amultiplied haversign signal; stripping off high frequencycharacteristics of the multiplied haversign signal to create a haversignsignal; receiving a first inductor current through a first switch;comparing a magnitude of the haversign signal with a value of the firstinductor current; and generating a first driving signal to turn off thefirst switch if the value of the first inductor current is larger thanthe magnitude of the haversign signal.
 7. The method of claim 6, furtherincluding receiving a second inductor current through a second switch;comparing a magnitude of the haversign signal with a value of the secondinductor current; and generating a second driving signal to turn off thesecond switch if the value of the second inductor current is larger thanthe magnitude of the haversign signal.
 8. The method of claim 7, furtherincluding, receiving the AC input voltage at a first inductor and asecond inductor; rectifying the AC input voltage using the first switch,the second switch, a first diode, and a second diode to produce anintermediate DC voltage.
 9. A method of power factor correction,comprising: receiving an AC input voltage; generating an intermediate DCvoltage; generating, at a waveform generator, a haversign waveform;generating a pulsed signal based on the intermediate DC voltage;multiplying the haversign waveform and the pulsed signal to create amultiplied haversign signal; stripping off high frequencycharacteristics of the multiplied haversign signal to create a haversignsignal; receiving a first inductor current through a first switch;comparing a clipping threshold value with a value of the first inductorcurrent; and generating a first driving signal to turn off the firstswitch if the value of the first inductor current is larger than themagnitude of the clipping threshold value.
 10. The method of claim 9,further including, receiving a second inductor current through a secondswitch; comparing the clipping threshold value with a value of thesecond inductor current; and generating a second driving signal to turnoff the second switch if the value of the second inductor current islarger than the clipping threshold value.
 11. The method of claim 10,further including, receiving the AC input voltage at a first inductorand a second inductor; rectifying the AC input voltage using the firstswitch, the second switch, a first diode, and a second diode to producean intermediate DC voltage.